Programmable Gain Trans-Impedance Amplifier Overload Recovery Circuit

ABSTRACT

Embodiments of an apparatus for measuring the leakage current of capacitive components is taught. One embodiment includes a first stage amplifier configured to receive an input from a serially-connected capacitive component at an inverting input and a feedback resistor in a feedback path of the first stage amplifier. A resistance value of the feedback resistor is programmable based on an expected value of the leakage current and a corresponding voltage output.

TECHNICAL FIELD

The present invention relates in general to a test for capacitivecomponents.

BACKGROUND

In a known apparatus and method for testing a capacitive component, suchas a capacitor, the component is first charged to a desired voltage.Then, the leakage current is measured. An out-of-range value for theleakage current can indicate that the component is faulty.

BRIEF SUMMARY

Embodiments of the invention provide a way to speed up testing ofcapacitive components, which is particularly desirable in automated,high-volume manufacturing processes. In particular, the inventionprovides embodiments of an apparatus that quickly recovers fromoverloads in order to perform the desired test.

The inventive features of certain embodiments are described in moredetail below.

BRIEF DESCRIPTION OF THE DRAWINGS

The description herein makes reference to the accompanying drawingswherein like reference numerals refer to like parts throughout theseveral views, and wherein:

FIG. 1 is a schematic diagram of a trans-impedance amplifier inaccordance with one embodiment of the invention;

FIG. 2 is a schematic diagram of a trans-impedance amplifier inaccordance with another embodiment of the invention;

FIG. 3 is a schematic diagram of a trans-impedance amplifier inaccordance with yet another embodiment of the invention;

FIG. 4 is a schematic diagram of a trans-impedance amplifier inaccordance with FIG. 3 illustrating a programmable gain of each stage;

FIG. 5 is plan view of an electronic component handling machine withwhich embodiments of the invention can be used; and

FIG. 6 is a schematic diagram of a fast recovery current sinkincorporated with the trans-impedance amplifier according to FIG. 1.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

When testing capacitive components such as multi-layer ceramiccapacitors (MLCCs) for leakage current, there exists a large currentthat the sensing circuitry endures due to the uncharged capacitivecomponent passing some of the charging current to the sense circuitry,where the majority of the passed current is sunk into charging diodes.Ideally, the capacitive component no longer passes current once fullycharged. In reality, leakage current passes, and the accuratemeasurement of this current is an important measure of the quality,specifically the insulation resistance, of the capacitive component.

Conventionally, such testing is performed using circuits including oneor more operational amplifiers (op amps) with a respective gain. For anop amp, there is a finite limitation on how large its input signal canbe in a given configuration before the op amp goes into saturation. Whenan op amp is in saturation, the output of the op amp is fixed at itspositive or negative supply voltage until the input signal is reduced toa point where the op amp is in its operating range and can provide acorrect output signal. How fast the op amp can come out of saturation isa function of several factors, including but not limited to its outputcurrent sourcing capability, configuration and loading. When a circuitcontains more than one op amp, an overload at the input of the firststage op amp causing the output to saturate could put the followingstages into a saturation condition as well.

When testing in data acquisition, and where speed is important, quickrecovery from overload conditions is desirable so that accurate data canbe obtained at a given time and/or so that the next set of data can bequickly obtained. Accordingly, embodiments of the invention describedherein actively drive the measuring circuitry out of such overloads.

FIG. 1 shows a trans-impedance amplifier according to one embodiment ofthe invention. FIG. 1 includes a current source 10 providing a constantcurrent Iin. The current source 10 can be implemented by any number ofcircuit designs within the knowledge of those in this field given thedesired output value. For example, the current source 10 can representthe combination of a voltage source supplying a variable voltage coupledto a programmable current source as described in commonly-assigned U.S.patent application Ser. No. 11/753,177, filed on May 24, 2007 andentitled Capacitive Measurements with Fast Recovery Current Return,which is incorporated herein in its entirety by reference.

In FIG. 1, a capacitive component to be tested, hereinafter device undertest (DUT) 20, is shown in phantom. As discussed in additional detailhereinafter, a number of such devices are sequentially tested uponconnection to the current source 10 and the measuring circuitryincluding the trans-impedance amplifier.

The DUT 20 is coupled in series to the current source 10, and thecurrent passing therethrough is supplied to the inverting input of an opamp arranged as a current sense amplifier 12 through a resistor 14. Thenon-inverting input of the current sense amplifier 12 is grounded.Feedback to the inverting input of the current sense amplifier 12 fromVout is provided by a resistor 16 in parallel with a capacitor 18. Thevalue of resistor 16 is Rf.

The transfer function of the current sense amplifier 12 is Vout=−Iin*Rf,which means that for a given input current, the output voltage Vout isthe mathematical product of the input current and the resistance valueRf. The ohmic value of Rf is selected based on design requirements. Forexample, given a measured leakage current is a value between +/−1 mAwith full scale representing +/−5V at Vout. A value of Rf of 5kΩ couldbe used. Full scale is defined by the designer based on application.Programming of the value of Rf is discussed in more detail hereinafterwith respect to FIG. 4.

FIG. 2 includes a trans-impedance amplifier with the current senseamplifier 12 of FIG. 1 as a first stage with the addition of anon-inverting gain stage in the form of an op amp arranged as anon-inverting voltage amplifier 22. One or more additional gain stages,such as that seen in FIG. 2 may be needed when one is trying to monitorlow level signals. In turn one could increase the value of Rf in thefirst amplifier. However, the inventors found that this solution can beproblematic if the op amp of the current sense amplifier 12 reached itslimits on gain or bandwidth. Also, in some circumstances, keeping theohmic value of Rf smaller can help with the noise gain of the system.The additional amplifier provided by the second stage non-invertingvoltage amplifier allows the application of the same logic to determineranges and full scale vales as for the first stage described previously.

In FIG. 2, the output of the first stage, voltage Vin, is provided tothe non-inverting input of the voltage amplifier 22. Feedback from theoutput of the voltage amplifier 22 to its inverting input is providedthrough resistor 24 having a resistance value of R3, and the invertinginput is grounded through resistor 26 having a resistance value of R2.The transfer function of this gain stage is Vout=Vin*(1+R3/R2). Asdescribed previously, and where the output of the first stage is Vin,the transfer function of the first stage is Yin=−Iin*Rf. Accordingly,with the known current Iin and a desired maximum input voltage Vin and adesired maximum output voltage Vout, one can assign appropriate valuesfor Rf, R3 and R2. For example, where the leakage current to be measuredis expected to fall between +/−1 μA (instead of +/−1 mA as in the aboveexample), and full scale represents +/−5V, the ohmic value of Rf isequal to 1.25 MΩ where the gain (1+R3/R2) of the second stage is equalto 4.

Although the second stage is shown as a non-inverting voltage amplifier22, the op amp can instead be arranged as a non-inverting unity gainbuffer as described in additional detail hereinafter with respect toFIG. 4. Also, more than one stage of amplification can be included ifdesired, based on the teachings herein.

FIG. 3 includes the first and second stages shown in FIG. 2 and adds anadditional feedback circuit from the output of the second stage to theinverting input of the first stage. As shown in FIG. 3, the output ofthe voltage amplifier 22 of the second, non-inverting gain stage isconnected to two back-to-back Zener diodes 28. The Zener diodes 28 areconnected in series to ground through a resistor 30. An op amp arrangedas a non-inverting buffer 32 follows the Zener diodes 28. That is, theZener diodes 28 are coupled to the non-inverting input of the buffer 32,and a feedback path 34 is provided between the output of the buffer 32and its inverting input. The output of the buffer 32 is connected toback-to-back signal diodes 36, which are in turn coupled to theinverting input of the current sense amplifier 12 of the first stage.More specifically, resistor 14 of FIGS. 1 and 2 is replaced by resistors14 a and 14 b in FIG. 3, and the feedback circuit provided by theback-to-back Zener diodes 28, non-inverting buffer 32 and back-to-backsignal diodes 36 is connected to the node 38 between resistors 14 a and14 b.

In this circuit, input currents in the desired, expected range allow thecircuit to behave in a linear fashion as determined by the transferfunctions of the stages as previously discussed. During normaloperation, the back-to-back Zener diodes 28 prevent the flow of currentto the feedback path because Vout is below the turn on voltage of theZener diodes. The non-inverting buffer 32 is inactive, and theback-to-back signal diodes 36 prevent the flow of current from thejunction 38 to the output of the non-inverting buffer 32. However, wherethe input currents into the circuit (that is, the input currents intothe inverting input of the first stage current sense amplifier 12) areout of range and large enough to drive the output of the amplifier toeither of its rails, the voltage output Vout from the voltage amplifier22 begins to move towards its power supply rails. Once large enough toturn on the Zener diodes 28, Vout will be at the clamping voltage of theZener diodes 28. Then, the clamping voltage is provided to thenon-inverting buffer 32, which is desirably selected for a high-currentdrive capacity. The current from the output of buffer 32 is supplied tothe signal diodes which begin to conduct after enough current frombuffer 32 is supplied to them, which in turn actively drives thepotential at node 38 lower. As this potential lowers, the input signalseen by the current sense amplifier 12 reduces to allow the amplifier tostart recovery from the overload at its input.

Although FIG. 3 illustrates a feedback circuit from the output of thesecond stage to the inverting input of the first stage, the feedbackcircuit of FIG. 3 could implemented with a single stage as taught inFIG. 1. This is generally less desirably at least for stability reasons,but it is possible. Also, the configuration of FIG. 3 could beimplemented with the non-inverting amplifier 22 of FIG. 2 being replacedwith a buffer as described with respect to FIG. 4.

The gains in the circuit topology can be programmable as shown in FIG.4. In FIG. 4, the gain of both the first stage and the second stage areprogrammable depending on the application. In addition to theconfiguration shown in FIG. 3, FIG. 4 includes a field-programmable gatearray (FPGA) 50 coupled to two feedback circuits for the current senseamplifier 12. More specifically, a first feedback circuit 52 includesresistor 16 a and capacitor 18 a arranged in parallel, and a secondfeedback circuit 54 includes resistor 16 b and capacitor 18 b arrangedin parallel. The value of either 16 a or 16 b is Rf, as previouslydescribed with respect to resistor 16. More than two feedback circuitsare possible. The FPGA 50 is programmed to switch in, that is, enableeither the first or second feedback circuit 52, 54 depending on ControlSignals 1 and 2. Similarly, optional control is provided for a switch 56coupled to the resistor 26 of the non-inverting voltage amplifier 22.Through the application of Control Signal 3 to switch 56, the resistor26 (with value R2) can be switched in and out of the circuitry. As thoseskilled in the art will recognize from this description, omittingresistor 26 would change the configuration of the op amp from that ofthe non-inverting voltage amplifier 22 to a non-inverting unity gainbuffer. This configuration is useful when the output signal Vin does notneed to be amplified. As in the discussion of FIG. 3, the second stagecan be omitted in its entirety if desired.

Control signals 1, 2 and 3 in this embodiment are provided by amicrocomputer including a random access memory (RAM), a read-only memory(ROM), keep alive memory (KAM), a central processing unit (CPU), etc.,in addition to various input and output connections. In the applicationdescribed below with respect to FIG. 5, for example, the microcomputeroperates a software program to perform the described testing, includingthe presentation of a set up menu by the user including information suchas the expected leakage current, which depends on the size of thecapacitive component, and the corresponding full scale voltage.Responsive to this menu, the microcomputer can program the gain throughthe Control Signals 1, 2 and 3 provided to the FPGA 50 and switch 56. Ofcourse, the functions of the FPGA 50 could be implemented by one or morehardware components. Any number of solid-state switches could be used toimplement the switch 56.

Having programmable gains in the circuit topology allows any metercontaining the circuitry to have a large dynamic range. For example, themeasurement capability could range from leakage current values frombetween +/−1 mA with resolution down to +/−200 pA. This task isdifficult to accomplish using only fixed values of Rf, R3 and R2 whensuch precision measurements are needed. Subdividing this range into aplurality of regions makes the task easier for the hardware involved.Within these ranges, ohmic values for Rf, R3 and R2 can be assigned bydefining a full scale voltage for each range and gain values for thesecond stage as described above. When a large dynamic range is neededfor the instrument, the ohmic value of Rf of current sense amplifier 12and the gain of amplifier 22 are desirably programmed. Then, independentof range, if the input current to the circuit is large enough to causean overload at current sense amplifier 12, driving the output ofamplifier 22 towards the Zener diode 28 turn on voltage. Once thatvoltage is reached, the feedback network activates, assisting thecircuit to return to its linear range much faster than amplifier 12 or22 could do by themselves and turning off once the product of the outputvoltage of current sense amplifier 12 and the gain of amplifier 22 isbelow the Zener diode turn on voltage. The circuit is back in its linearrange.

As mentioned, when a range of leakage current values is expected,subdividing the ranges and controlling the programmable gain based oneach range is desirable. The following table provides values for fourdifferent ranges of leakage currents as an example of the application ofthe programmable gain teachings of the invention.

Leakage current Value Rf Gain of range (+/−) (kΩ) second stage 3 μA 3244 12 μA 324 1 50 μA 20 4 200 μA 20 1

These ranges can be implemented in the embodiment of FIG. 4 byassociating the value RF of 324 kΩ with resistor 16 a and associatingthe value Rf of 20 kΩ with resistor 16 b. These values could then beselectively switched by the FPGA 50. The resistance values of resistors24 and 26 would be set such that the gain is 4, for example, the valueR3 could be 3 kΩ, while the value of R2 could be 1 kΩ. The gain ofamplifier 22 could then be selectively switched by switch 56 to changethe gain between 1 and 4 depending on the desired range.

The circuitry according to any of FIGS. 1-4 can be implemented as partof a stand-alone test device in any number of applications for testingcapacitive components. The trans-impedance amplifier can also beimplemented in a separate device from the current source. Then, thecurrent source can be any programmable computer-controlled source, suchas models of the 54XX power supply available from Electra ScientificIndustries, Inc. of Portland, Oreg., the assignee of the present patentapplication. One particularly desirable use of the trans-impedanceamplifier is in electronic component handlers that test a high volume ofelectronic components in a relatively short period of time. Thesehandlers include but are not limited to products sold by ElectroScientific Industries, Inc., which sells a variety of electroniccomponent handlers including but not limited to a high volume MLCCtester sold as the Model 3500.

One electronic component handling machine is illustrated incommonly-assigned U.S. Pat. No. 5,842,579 entitled Electrical CircuitComponent Handler, which is incorporated herein in its entirety byreference. FIG. 5 shows an overall pictorial view of the electricalcircuit component handler 100. The handler 100 has a loading frame 112defining a loading zone 130, a plurality of test modules 114 defining atest zone 115 and a blow-off 160 defining a blow off zone 170. Inoperation, electronic components are passed through loading frame 112 inloading zone 130 to be individually drawn into test seats 124 found on atest plate (not shown in its entirety) with the assistance of a vacuum.After testing a component in the test zone 115, the component moves tothe blow off zone 170, where the blow-off 160 removes the vacuum andsorts the part based on the results of the test(s).

Although not shown in detail, an embodiment of the invention and thecurrent source 10, if separately implemented, would be electricallycoupled to the test modules 114 for testing each component in the testzone 115. Namely, the components in the test seats 124 are subjected toa number of tests in the test zone 115 through the use of the testmodules 114. For example, when MLCCs are tested, data is generallyprovided on, for example, the capacitance, dissipation factor andinsulation resistance. The data obtained from testing can then be usedto sort the parts by tolerance and find those parts that are defective.

As explained briefly above, in operation, when an uncharged capacitorenters for testing, here being placed in a test seat 124, a chargecurrent and charge voltage are applied to this DUT 20 to sequentiallyperform tests in sequence according to individual manufacturerrequirements. With respect to the insulation resistance (IR) test, theapplication of the charge current from current source 10 to the DUT 20results in a large current, producing an overload in the trans-impedanceamplifier, starting with an overload of the current sense amplifier 10.As the DUT 20 charges, the current seen at the inverting input of thecurrent sense amplifier 12 begins to lower. After a short time, theinput to the current sense amplifier 12 is the leakage current. Themeasured leakage current represents the insulation resistance of the DUT20. The recovery time for the current sense amplifier 12 and, whereapplicable, the voltage amplifier 22, is minimized by appropriate gainselection for each stage and the feedback circuit of FIG. 3 when used. Asmall recovery time is especially important when the DUT 20 is a largevalue capacitor, where the leakage current is relatively high. If themeasurement is taken before recovery is completed, that is, before theend of the overload(s), the measurement may capture the response of themeasurement circuitry recovery and not the actual leakage current. Inhigh-speed testing, this is more of a risk.

The problems of initial overload and the resulting recovery time canalso be complicated by other tests that may be performed before theinsulation resistance (IR) test, One such test is a contact check test.The contact check test is used to verify that the part to be tested,such as DUT 20, has properly arrived at the test station. According toone known implementation, a 1 volt peak-to-peak, high frequency sinewave is generated through the DUT 20, and the resultant AC current ismeasured and compared to a predetermined threshold that indicates thepresence of the DUT 20. Sense circuitry separate from thetrans-impedance amplifier according to embodiments of the presentinvention perform this measurement from a common entry point. Thisapplication of the charge voltage can cause the circuitry of any ofFIGS. 1-3 to overload. Depending on how much soak time is allowed beforethe IR measurement is taken, fast recovery of the current senseamplifier 12 and the voltage amplifier 22 may be needed due to thisearlier test.

As can be seen, the circuitry described in the present applicationprovides an alternative solution to similar problems described in U.S.patent application Ser. No. 11/753,177. However, the inventive conceptstaught therein can be incorporated with the teachings of the presentinvention to obtain additional benefits. FIG. 6 illustrates the optionaladdition to the measurement circuitry of FIG. 1 of a fast recoverycurrent sink 46 in accordance with the teachings of U.S. patentapplication Ser. No. 11/753,177.

In FIG. 6, the fast recovery current sink 46 is connected at a tap 60between the DUT 20 and resistor 14 and is grounded. A diode clamp 40 isprovided by two diodes 42, 44 arranged in parallel such that the anodeof the first diode 42 and the cathode of the second diode 44 aregrounded and the cathode of the first diode 42 and the anode of thesecond diode 44 are electrically coupled to the tap 60. The clamp diodes42, 44 provide a current return path for the charge current of the DUT20. In addition, the clamp diodes 42, 44 provide input protection forthe current sense amplifier 12 if the DUT 20 short circuits by clampingthe voltage of the current source 10. Note that the clamp diodes 42, 44can be similarly incorporated in each embodiment of the invention,although they are not shown in FIGS. 1-4 to simplify the description ofthe trans-impedance amplifier.

The switch 70 of U.S. patent application Ser. No. 11/753,177 that isconnected in parallel across the diode clamp 40 is implemented here by asolid-state relay, by example only a PVG612 power MOSFET photovoltaicrelay available from International Rectifier of El Segundo, Calif. Acontrol signal 48 from a controller closes the switch 70 while the DUT20 is being charged by the current source 10. Then, the switch 70 isopened before the leakage current is measured by the trans-impedanceamplifier.

Using the teachings of the present invention, the need to coordinatetiming between switches enabling and disabling the charging circuitry(including the sink 46) and the measurement circuitry described in U.S.patent application Ser. No. 11/753,177 is eliminated. As the currentsink 46 is switched open before leakage current is measured, a change inthe potential of the current sense amplifier 12 results. Accordingly,there is a resulting change at the output of the current sense amplifier12 and the voltage amplifier 22, where used. The overload recoverycomponents of the trans-impedance amplifier taught herein then assist inreducing any voltage disturbance, allowing the leakage currentmeasurement to be taken quickly after the current sink 46 is switched.

The combination of these teachings is particularly useful when testinghigh value capacitors. This is because when the current sink 46 isswitched out, there is an immediate change in the magnitude of the inputto the current sense amplifier 12 due to the high-impedance of theentire trans-impedance amplifier. This magnitude depends on the inputimpedance value Rin of the trans-impedance amplifier and the value Rf ofthe current sense amplifier 12. The voltage gain of the trans-impedanceamplifier follows the transfer function Vout=−Rf/Rin. If the value Rf isas large as is needed for nano-amp and pico-amp measurements, an initialoverload to the current sense amplifier 12 is likely. Due to theoverload recovery of the trans-impedance amplifier, the overload quicklyresolves.

Although FIG. 6 shows the fast recovery current sink 46 coupled to thetrans-impedance amplifier of FIG. 1, the fast recovery current sink 46can be incorporated into any of the embodiments thereof, including thoseshown and described with respect to FIGS. 2-4.

Capacitive component leakage current measurements in productionenvironments require both accuracy and speed. Embodiments of atrans-impedance amplifier with overload recovery allow for a quickreduction in any initial overload in the front end circuitry, thuslimiting the delay in the measurement of the leakage current past thetime the capacitive component being tested is fully charged. Moreover,implementing an embodiment of the trans-impedance amplifier reduces therisk that the measurement so taken will reflect its recovery instead ofthe actual leakage current.

The above-described embodiments have been described in order to alloweasy understanding of the present invention, and do not limit thepresent invention. On the contrary, the invention is intended to covervarious modifications and equivalent arrangements included within thespirit and scope of the appended claims, which scope is to be accordedthe broadest interpretation so as to encompass all such modificationsand equivalent structures as is permitted under the law.

1. An apparatus for measuring a leakage current of a capacitive component, the apparatus comprising: a first stage amplifier configured to receive an input from a serially-connected capacitive component at an inverting input; and a feedback path of the first stage amplifier coupled between an output of the first stage amplifier and the inverting input, a resistance value of the feedback path dependent on an expected value of the leakage current and a corresponding full scale voltage of the first stage amplifier.
 2. The apparatus according to claim 1 wherein the resistance value of the feedback path is programmable, the apparatus further comprising: means for changing the resistance value to one of a plurality of values.
 3. The apparatus according to claim 2, further comprising: a second stage amplifier coupled to the output of the first stage amplifier, the second stage amplifier including a programmable gain.
 4. The apparatus according to claim 3 wherein the programmable gain is switchable from unity gain to a value greater than unity gain.
 5. The apparatus according to claim 1, further comprising: a second stage amplifier coupled to the output of the first stage amplifier.
 6. The apparatus according to claim 5 wherein the output of the first stage amplifier is coupled to a non-inverting input of the second stage amplifier.
 7. The apparatus according to claim 6 wherein the second stage amplifier includes a programmable gain, the apparatus further comprising: a switch configured to switch the programmable gain from unity gain to a value greater than unity gain.
 8. The apparatus according to claim 7, further comprising: a feedback path from an output of the second stage amplifier to a summing junction at the inverting input of the first stage amplifier.
 9. The apparatus according to claim 8 wherein the feedback path from the output of the second stage amplifier to the summing junction at the inverting input of the first stage amplifier comprises: means for preventing a flow of current from an output of the second stage amplifier to the summing junction at the inverting input of the first stage amplifier until the output of the second stage exceeds a defined range of values; and means for limiting a flow of current in a direction from the summing junction to the output of the second stage amplifier.
 10. The apparatus according to claim 9 wherein the defined range includes values greater than a saturation voltage of the second stage amplifier.
 11. The apparatus according to claim 8 wherein the feedback path comprises: back-to-back Zener diodes coupled to the output of the second stage amplifier; a buffer amplifier including a non-inverting input coupled to the back-to-back Zener diodes; and back-to-back signal diodes coupled to an output of the buffer amplifier and to the summing junction of the inverting input of the first stage amplifier.
 12. The apparatus according to claim 1, further comprising: a second feedback path from the output of the first stage amplifier to the inverting input of the first stage amplifier, the second feedback path configured to reduce an input to the inverting input when an output value of the first stage amplifier indicates saturation of the first stage amplifier.
 13. The apparatus according to claim 1 wherein the resistance value of the feedback path is provided by a programmable feedback resistor.
 14. The apparatus according to claim 13, further comprising: a programmable device configured to switch the programmable feedback resistor from a first value to a second value dependent on the expected value of the leakage current and the corresponding full scale voltage of the first stage amplifier.
 15. The apparatus according to claim 14, further comprising: a second stage amplifier coupled to an output of the first stage amplifier and having a programmable gain; and a switch configured to switch the programmable gain of the second stage amplifier from unity gain to a value greater than unity gain.
 16. The apparatus according to claim 13, further comprising: a second stage amplifier coupled to an output of the first stage amplifier and having a programmable gain; a second feedback path from an output of the second stage amplifier to the inverting input of the first stage amplifier, the second feedback path configured to reduce an input to the inverting input when an output value of the second stage amplifier indicates saturation of the second stage amplifier; and a switch configured to switch the programmable gain of the second stage amplifier from unity gain to a value greater than unity gain.
 17. The apparatus according to claim 16 wherein the second feedback path comprises: means for preventing a flow of current from the output of the second stage amplifier to the inverting input of the first stage amplifier until the output value of the second stage amplifier indicates saturation of the second stage amplifier; and means for limiting a flow of current in a direction from the inverting input of the first stage amplifier to the output of the second stage amplifier.
 18. The apparatus according to claim 16 wherein the second feedback path comprises: back-to-back Zener diodes coupled to the output of the second stage amplifier; a buffer amplifier including a non-inverting input coupled to the back-to-back Zener diodes; and back-to-back signal diodes coupled to an output of the buffer amplifier and to the inverting input of the first stage amplifier.
 19. (canceled)
 20. (canceled)
 21. The apparatus according to claim 1, further comprising: an input resistor coupled to the inverting input of the first stage amplifier and configured to receive the input from the serially-connected capacitive component.
 22. The apparatus according to claim 1 wherein the non-inverting input of the first stage amplifier is directly coupled to ground. 